Method and apparatus for timing synchronization at sub-sampled rate for sub-sampled wideband systems

ABSTRACT

A method of timing synchronization in sub-band based ultra wideband systems, includes obtaining a coarse estimate of an offset in a time domain at a sub-sampled rate, and obtaining a fine estimate of the offset in an analog domain. The method further includes correcting a timing in the analog domain by transforming the fine estimate to an equivalent phase for the correcting.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit under 35 USC 119(a) of Indian PatentApplication No. 189/CHE/2013, filed on Jan. 15, 2013, in the IndianPatent Office, and Korean Patent Application No. 10-2013-0035705, filedon Apr. 2, 2013, in the Korean Intellectual Property Office, the entiredisclosures of which are incorporated herein by reference for allpurposes.

BACKGROUND

1. Field

The following description relates to a method and an apparatus fortiming synchronization at a sub-sampled rate for sub-sampled widebandsystems.

2. Description of Related Art

Wideband communications are gaining importance for future wirelesscommunications because of their potential to support an extremely highdata rate of a gigabit per second. Power consumption is a critical issuein wideband systems considering a wide bandwidth involved. Developinglow power, low cost, and low interference wideband transceivers has ahuge commercial demand. A concept of sub-banding is being developed forwideband systems. With this concept, power saving is achieved in ultrawideband (UWB) systems.

Several technologies based on personal area networks (PANs) use thisband to build applications that can achieve medium to high datacommunication rates. In a method of sub-banding, a given bandwidth of500 MHz (or more) is divided equally into N sub-bands. These N sub-bandscan be used to provide multiple users access to channel bandwidth, toincrease rates of data communication by using different sub-bands totransmit different data streams, and improving communication performanceby using different sub-bands to transmit a same data stream.

In sub-band ultra wide band (S-UWB) systems, a transmitting deviceincludes a plurality of sub-band signal generators that generate aplurality of sub-band signals based on determined parameters, where eachof the sub-band signals includes modulated bit streams spread usingspreading codes. Further, at a receiver side, a receiving device of theS-UWB systems includes an analogue front end that receives an S-UWBsignal including the sub-band signals from the transmitting device overa UWB channel. The receiving device also includes a sampler that samplesthe S-UWB signal at a rate of a sub-band bandwidth. A saving in energyis attributed to base-band processing at the sub-sampling rate, whichmay obviate a need of a higher sampling rate ADC used in full-bandsystems.

SUMMARY

This Summary is provided to introduce a selection of concepts in asimplified form that are further described below in the DetailedDescription. This Summary is not intended to identify key features oressential features of the claimed subject matter, nor is it intended tobe used as an aid in determining the scope of the claimed subjectmatter.

In one general aspect, a method of timing synchronization in sub-bandbased ultra wideband systems, includes obtaining a coarse estimate of anoffset in a time domain at a sub-sampled rate, and obtaining a fineestimate of the offset in an analog domain. The method further includescorrecting a timing in the analog domain by transforming the fineestimate to an equivalent phase for the correcting.

The obtaining of the coarse estimate may include performing crosscorrelation of a sub-sampled signal with a sub-sampled training sequencein a digital domain.

The method may further include obtaining the coarse estimate for eachsub-band based on a frequency diversity order of a user.

The method may further include obtaining a final coarse estimate of theoffset from each sub-band by determining a maximum of estimates for acorresponding sub-band.

The obtaining of the fine estimate may include obtaining the fineestimate in the analog domain based on the final coarse estimate.

The obtaining of the fine estimate may include performing crosscorrelation of a delayed signal with a training sequence designed with asilence period.

A length of the training sequence may be equal to a number of orthogonalsubcarriers present over a bandwidth of the delayed signal.

The training sequence may be an up-sampled version of a base trainingsequence.

The training sequence of a sub-band may be orthogonal up to a lag gap ofan integer, the lag gap indicating a shift of sub-bands by a value ofthe integer.

The training sequence may be followed by the silence period of at leastfour orthogonal frequency division multiplexing symbols.

In another general aspect, an apparatus configured to perform timingsynchronization in sub-band based ultra wideband systems, includes acoarse estimation unit configured to obtain a coarse estimate of anoffset in a time domain at a sub-sampled rate, and a fine estimationunit configured to obtain a fine estimate of the offset in an analogdomain. The apparatus further includes a timing correction unitconfigured to correct a timing in the analog domain by transforming thefine estimate to an equivalent phase for the correction.

The coarse estimation unit may be configured to perform crosscorrelation of a sub-sampled signal with a sub-sampled training sequencein a digital domain.

The fine estimation unit may be configured to perform cross correlationof a delayed signal with a training sequence designed with a silenceperiod.

The apparatus may further include a phase locked loop circuit configuredto determine the equivalent phase based on the fine estimate, ananalog-to-digital converter configured to convert an input signal in theanalog domain to a sub-sampled signal in a digital domain to be used toobtain the coarse estimate, and an analog delay circuit configured todelay the input signal, the delayed input signal being used to obtainthe fine estimate.

In still another general aspect, a receiver configured to perform timingsynchronization in sub-band based ultra wideband systems, includes aprocessor configured to determine a first estimate of a timing offset ofan input signal, in a digital domain, and determine a second estimate ofthe timing offset in an analog domain based on the first estimate. Theprocessor is further configured to determine an equivalent phase to beused to correct a timing of the input signal, based on the secondestimate.

The processor may be configured to sub-sample the input signal togenerate a sub-sampled signal in the digital domain, and perform crosscorrelation of the sub-sampled signal with a sub-sampled trainingsequence in the digital domain to generate the first estimate.

The processor may be further configured to scale the first estimate, anddetermine the second estimate based on the scaled first estimate.

The processor may be configured to delay the input signal, and performcross correlation of the delayed input signal with a training sequencefollowed by a silence period, based on the first estimate, to generate acorrelated output.

The processor may be configured to determine the second estimate to bethe first estimate in response to the correlated output being greaterthan or equal to a predetermined threshold.

The processor may be configured to control a switching of ananalog-to-converter clock from an initial phase to the equivalent phasebased on a predetermined timing delay, to correct the timing of theinput signal.

Other features and aspects will be apparent from the following detaileddescription, the drawings, and the claims.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating an example of an apparatus thatperforms timing synchronization.

FIG. 2 is a block diagram illustrating an example of an apparatus thatperforms fine timing synchronization.

FIG. 3 is a circuit diagram illustrating an example of an analog delaycircuit for fine timing synchronization.

FIG. 4 is a graph illustrating an example of a comparison of a bit errorrate (BER) and a signal-to-noise ratio (SNR) of an orthogonal frequencydivision multiplexing (OFDM) based sub-band ultra-wide band (S-UWB)system with timing synchronization at 100 MHz and 1 GHz over an UWBchannel.

FIG. 5 is a graph illustrating another example of a comparison of a BERand a SNR of an OFDM-based S-UWB system with timing synchronization at 1GHz and 100 MHz over an UWB channel.

FIG. 6 is a diagram illustrating an example of a method of designing atraining sequence for timing synchronization at a sub-sampled bandwidth.

FIG. 7 is a schematic diagram illustrating an example of a frame formatof an OFDM-based S-UWB system.

Throughout the drawings and the detailed description, unless otherwisedescribed or provided, the same drawing reference numerals will beunderstood to refer to the same elements, features, and structures. Thedrawings may not be to scale, and the relative size, proportions, anddepiction of elements in the drawings may be exaggerated for clarity,illustration, and convenience.

DETAILED DESCRIPTION

The following detailed description is provided to assist the reader ingaining a comprehensive understanding of the methods, apparatuses,and/or systems described herein. However, various changes,modifications, and equivalents of the systems, apparatuses and/ormethods described herein will be apparent to one of ordinary skill inthe art. The progression of processing steps and/or operations describedis an example; however, the sequence of and/or operations is not limitedto that set forth herein and may be changed as is known in the art, withthe exception of steps and/or operations necessarily occurring in acertain order. Also, descriptions of functions and constructions thatare well known to one of ordinary skill in the art may be omitted forincreased clarity and conciseness.

The features described herein may be embodied in different forms, andare not to be construed as being limited to the examples describedherein. Rather, the examples described herein have been provided so thatthis disclosure will be thorough and complete, and will convey the fullscope of the disclosure to one of ordinary skill in the art.

Examples of a method and an apparatus for timing synchronization at asub-sampled rate for sub-sampled ultra wideband systems (UWB) aredescribed herein. The method is of, and the apparatus includes, asynchronizer with a high synchronization probability to obtain a betterchannel estimate.

The method of the timing synchronization includes a mixed mode design.Initially, estimation of timing information is carried out in a digitaldomain at a sub-sampled rate (100 MHz) after an analog-to-digitalconversion (ADC), and compensation is carried out over a real timeanalog signal of 500 MHz. Further, the estimated timing information isfed back to a fractional phase locked loop (PLL) to control a phase of asampling clock of the ADC which results in correction of a samplingtime. The method of the timing synchronization needs a suitable trainingsequence that takes care of a rate mismatch between the estimation andthe compensation in the digital domain and an analog domainrespectively. For example, the training sequence may be band-limited toa sub-sampled bandwidth.

FIG. 1 is a block diagram illustrating an example of an apparatus 100that performs timing synchronization. The timing synchronizationincludes two stages. In a first stage, the apparatus 100 estimates anoffset, and in a second stage, the apparatus 100 compensates a signalwith the estimated offset. That is, the apparatus 100 that performs thetiming synchronization includes a mixed mode design, as describedherein.

The estimation of the offset is performed in a digital domain at asub-sampled rate (100 MHz) after analog-to-digital converter (ADC), andthe compensation is performed over a real time analog signal of 500 MHz.The mixed mode estimation and compensation include processing in bothdigital and analog domains. Hence, the timing synchronization includesthe processing at a full rate and the sub-sampled rate. It is alsoassumed that a receiver system (e.g., the apparatus 100) is free fromfrequency offset errors.

The mixed mode design of the timing synchronization is based on atraining sequence pattern, where silent periods are added for othersub-bands. Further, training sequences for the timing synchronizationare added for a desired band in a preamble section dedicated forfrequency synchronization of a sequence of a frame. The trainingsequences for different sub-bands are protected from overlapping witheach other because of an aliasing effect of the sub-sampling.

Referring to FIG. 1, the apparatus 100 includes a coarse estimation unit101, a fine estimation unit 102, and a correction unit 103. The coarseestimation unit 101 performs coarse estimation of a timing offset ineach sub-band individually at the sub-sampled rate. The fine estimationunit 102 performs fine estimation of the timing offset at the full ratebased on the coarsely-estimated timing offset and an analog (i.e., radiofrequency (RF)) signal received in the analog domain, to finallygenerate an equivalent parameter corresponding to an estimated timingerror input. The correction unit 103 compensates the analog signal inthe analog domain based on the equivalent parameter.

The coarse estimation of the timing offset in each sub-band individuallyat the sub-sampled rate is described herein. Initially, the coarseestimation is performed in a sub-sampled domain. Consider the followingparameters as described herein.

Let S denote a modulated sequence transmitted over a time-dispersive UWBfading channel given as:

S={ . . . s(n),s(n+1), . . . }  (1)

Let h_(b) be complex baseband equivalent UWB channel coefficientsapplicable to a band b with an order P given as:

h _(b) ={h _(b)(1),h _(b)(2), . . . ,h _(b)(P)}  (2)

Let {tilde over (Y)} represent a corresponding received complex samplesequence at a UWB receiver front end shown as:

{tilde over (Y)}={ . . . y(n),y(n+1), . . . }  (3)

Further, N_(s) is a total number of samples in each OFDM symbol in atransmitted preamble. Then, sub-sampling of an n^(th) received sample ofan l^(th) OFDM symbol duration is expressed as:

$\begin{matrix}{{{y\left( {l,n} \right)} = {{\sum\limits_{b = 1}^{D}{\sum\limits_{p = 1}^{P}{{s\left( {l,{n - p - q_{b}}} \right)}{h_{b}(p)}}}} + {w\left( {l,n} \right)}}}{{0 < n < N_{s}},{0 < l < L},{1 \leq p \leq P}}} & (4)\end{matrix}$

In Equation 4, w(l, n) is a complex additive white Gaussian noise (AWGN)of the n^(th) sample of the l^(th) OFDM symbol, D is a total number ofsub-bands, L is a number of training symbols of a length N_(s) dedicatedfor the timing synchronization, and q_(b) is a timing offset of a b^(th)sub-band channel. The timing offset q_(b) is different for a differentsub-band channel response.

Assuming that packet detection is over, the timing synchronizationtargets to estimate a start of a first frequency synchronizationsequence of the preamble especially a start of a Fast Fourier transform(FFT) window for it.

Equation 4 indicates that each sub-band signal experiences a differentamount of a shift in the FFT window based on the corresponding timingdelay q_(b) contributed by the sub-band channel. But, due to thesub-sampling, the received sample y(l, n) experiences some equivalentoffset effect of all of the sub-bands.

An estimation of a timing offset of a full band channel is from theestimation of the individual sub-band timing offsets at the sub-sampledrate. This needs a proper design of a training sequence to avoidoverlapping of the sub-bands in the sub-sampled domain. This isimplemented by a silence period based training design. Hence, for atraining part of the preamble, for a first sub-sampled OFDM symboltransmitted over a sub-band 1, Equation 4 can be modified to:

$\begin{matrix}{{y\left( {1,n} \right)} = {{\sum\limits_{p = 1}^{P}{{s\left( {l,{n - p - q_{1}}} \right)}{h_{1}(p)}}} + {w\left( {l,n} \right)}}} & (5)\end{matrix}$

In the digital domain, received samples of Equation 5 arecross-correlated with known training sequences, i.e., a preamble pattern{t}={t(1),t(2), . . . , t(N)} that is band-limited to a sub-bandbandwidth. A cross-correlated output C_(b) (l, n) of the n^(th) sampleof the l^(th) OFDM symbol corresponding to the sub-band ‘b’ is obtainedas:

$\begin{matrix}{{{C_{b}\left( {l,n} \right)} = {\sum\limits_{i = 1}^{N_{s}}{{y\left( {l,{n + i}} \right)}{t^{*}(i)}}}}{for}{{1 \leq l \leq {L\; 1} \leq n \leq N_{s}};}{b \in \left\lbrack {1,{\ldots \mspace{14mu} D}} \right\rbrack}} & (6)\end{matrix}$

In Equation 6, * denotes a complex conjugate operation. Thecoarsely-estimated timing offset {circumflex over (τ)} is obtained byfinding a maximum delay point out of correlation peaks for all thesub-bands, using Equation 7 below:

$\begin{matrix}{{{\hat{\tau} = {\max \left( {\max \left( \frac{C_{b}}{{var}\left( C_{b} \right)} \right)} \right)}};}{b \in \left\lbrack {1,{\ldots \mspace{14mu} D}} \right\rbrack}} & (7)\end{matrix}$

The above Equation 7 determines an instant of a maximum delayedmultipath with significant energy of the full band channel as viewed bya sub-sampled window. A major portion of the full band channel energylies within an equivalent scaled-up version α{circumflex over (τ)}(α=sub−sampling factor=N) of the coarsely-estimated timing offset.

The fine estimation of the timing offset is described herein. The fineestimation unit 102 estimates the timing offset at a higher samplingrate in the analog domain. The fine estimation unit 102 receives thescaled coarsely-estimated timing offset in the digital domain, and thedelayed analog signal from an analog delay circuit. The analog delaycircuit delays the analog signal to introduce a delay due to the coarseestimation, the scaling up of the coarsely-estimated timing offset, andlatency in a feedback path. The delayed analog signal is stored in abuffer. The fine estimation is demonstrated in FIG. 2.

Once the coarse estimation and the fine estimation are done, thecorrection unit 103 performs a timing correction based on thefinely-estimated timing offset {circumflex over (τ)}_(final) inoperations as described herein. The timing correction is performed inthe analog domain at a higher rate, once a rest part of the preamblestarts arriving. The timing correction is performed by changing asampling clock phase of the ADC to the equivalent phase, which resultsin changing a sampling time for a next part of the frame.

The finely-estimated timing offset is fed to a fractional PLL of thecorrection unit 103, and the fractional PLL generates an equivalentphase {circumflex over (θ)} based on the finely-estimated timing offset.A timing buffer of the correction unit 103 controls a switching of anADC clock from an initial phase to the generated equivalent phase basedon a predefined timing delay. A flow of the analog signal in theapparatus 100 during the coarse and fine estimation and the timingcorrection is controlled by two switches. In FIG. 1, the switches are inposition 1 during an estimation phase.

The switches are connected to the position 1 until thecoarsely-estimated timing offset, the finely-estimated timing offset,and the equivalent phase are estimated. Then, both of the switches aremoved to position 2 to compensate based on the equivalent phase, and tofeed the sampled analog signal to an FFT after the sub-sampling.

In an example, the timing synchronization can also be achieved byobtaining the coarsely-estimated and finely-estimated timing offsetsfully in the digital domain. There exists applications of sub-sampled,orthogonal frequency division multiplexing (OFDM) based S-UWB systems,where the synchronization can be performed only in the digital domain,avoiding the complexity of the analog domain. The timing synchronizationin the digital domain is described herein with the following examples.

In an example, where Quality of Service (QoS) is achieved by improvisingcode diversity instead of frequency, the same data of a user is sentover different orthogonal codes in a single sub-band to achieve adiversity gain and, hence, improved QoS (for example, bit error rate(BER)) performance. Sub-bands are used to enhance a data rate for oneuser or to support multiple users in a network. In this example, allsignals in a sub-band will experience the same timing offset estimatedin a coarse estimation and, hence, the same amount of shift in an FFTwindow.

This coarse estimation can be improved by putting a predefined thresholdon a correlation output. The predefined threshold is a function of aroot mean square (RMS) delay spread of the full band channel. Correctioncan be carried out in the sub-sampled digital domain by shifting asequence by an estimated instant. This method eliminates the need offine estimating of a timing offset and timing correction. This methodalso provides fast estimation compared to a mixed mode solution fortiming synchronization.

If signals are transmitted with a diversity order of 1 along sub-bands,the signals do not add up automatically after de-spreading by orthogonalcodes. The orthogonal codes cancel an effect of interference of allother codes allotted to unwanted sub-bands. Similarly, for a diversityorder of 2, along a frequency axis, fine estimation can be completelyperformed in the digital domain as per the method described in the aboveexamples, and compensation can be performed in the analog domain.

Further, in another example, where parallel processing for a diversityalong a frequency, a digital solution is provided by finely estimating atiming offset, and providing timing correction, channel estimation andcorrection individually for each sub-band signal. Then, the signals ofthe sub-bands involved to achieve diversity are added after de-spreadingand fed to a demodulator. This process increases complexity, which isproportional to a diversity order.

FIG. 2 is a block diagram illustrating an example of an apparatus 200that performs fine timing synchronization. A delayed analog signal iscross-correlated with an analog template of the same training sequenceutilized for coarse estimation of a timing offset in a digital domain,after properly scaling up the coarsely-estimated timing offset. Thecross-correlation in an analog domain is performed in two operations asdetailed below.

Referring to FIG. 2, the apparatus 200 includes an analog delay circuit201, a multiplier 202, an analog integrator 203, a decision unit 204,and a PLL circuit 205. Initially, the analog delay circuit 201 receivesa buffered analog signal and feedback from the digital domain, andprovides a delay resolution of 1/B (B=full bandwidth) to delay theanalog signal. For a channel bandwidth of 500 MHz, a 2 nsec delayresolution may be used.

The multiplier 202 receives the delayed analog signal and a trainingsequence, and multiplies the delayed analog signal by the trainingsequence. The analog integrator 203 receives an output of the multiplier202, and cross-correlates the output to generate a correlated output as:

$\begin{matrix}{{{C^{\prime}(\kappa)} = {\int_{t = \kappa}^{T + \kappa}{{\overset{\sim}{s}\left( {t - \kappa} \right)}{{\overset{\sim}{t}}_{r}(t)}{t}}}},{1 \leq \kappa \leq {\alpha \hat{\tau}}}} & (8)\end{matrix}$

For low delay spread channels, a normalized correlation peak gives afinely-estimated timing offset. For higher delay spread channels,initially the correlation peak is utilized to correct a timing error ofthe incoming signal.

Referring to FIG. 1, after channel estimation in the digital domain, anestimation of a channel delay spread is performed and fed back to athreshold selector of the fine estimation unit 102. As per the delayspread of different UWB channel models, the threshold selector selectsand outputs a predefined threshold (λ).

Referring again to FIG. 2, the threshold selector output (i.e., thepredefined threshold) is fed back to the decision unit 204 in the analogdomain. The decision unit 204 determines the finely-estimated timingoffset {circumflex over (τ)}_(final) by comparing the correlated outputC′(κ) with the threshold. {circumflex over (τ)}_(final) represents aninstant when C′(κ)≧λ, and the decision unit 204 outputs thefinely-estimated timing offset to the PLL circuit 205, e.g., of thetiming correction unit 103 of FIG. 1.

FIG. 3 is a circuit diagram illustrating an example of an analog delaycircuit for fine timing synchronization. The compact circuit shown inFIG. 3 provides a controllable analog delay to an input analog signalV_(i) that can be accurately predicted if an operational amplifier(OPAMP) (e.g., CLC428) with a wide bandwidth is selected. With a givenvalue of a resistor R (95.3 ohms), a capacitor C (63 picofarads (pF)),and other resistors R_(g) and R_(f) (249 ohms), a delay of approximately(63×10⁻¹²)*(95.3×2), which equals to 12 nsec, can be generated andapplied to the input analog signal V_(i) to generate an output analogsignal V_(o). This delay can be tuned to 2 nsec. OPAMP integratorscapable of handling a 500 MHz wide signal may also possible by usingintegrated circuits (ICs), such as AD 8045 and AD 8099.

FIG. 4 is a graph illustrating an example of a comparison of a BER and asignal-to-noise ratio (SNR) of an OFDM-based S-UWB system with timingsynchronization at 100 MHz and 1 GHz over an UWB channel. The BER of thesystem with the timing synchronization for a variable diversity orderranging from 1 to 5 is shown in FIG. 4.

A performance of the described timing synchronization apparatus andmethod is evaluated through a simulation of the BER of the OFDM-basedSUWB system. Simulation parameters considered for this simulationanalysis is tabulated in Tablet. The performance of a 100 MHzsynchronizer is also compared with a 1 GHz synchronizer to calculate anestimation loss in a sub-sampled domain. The performance is obtainedover the IEEE 802.15.4 channel model CM3, which is considered to be alow delay spread channel with an RMS delay spread of 11 nsec.

TABLE 1 Simulation Parameters Sr. No Simulation Parameter Value/Type 1UWB Bandwidth (B) 500 MHz 2. Number of Sub-bands (N)  5 3 Sub-samplingrate 100 MHz 4 Chip duration/sampling time  10 ns 5 FFT size 32 6 CPlength 10 7 OFDM symbol duration 420 ns 8 Data subcarriers 32 9Spreading Codes Walsh Hadamard (WH) 10 Spreading code length (P)  8 11Modulation Type BPSK 12 Coding Uncoded 13 Payload 350 OFDM symbols/frame14 Data Rate 54 Mbps (with diversity 5) 15 Channel IEEE 802.15.4,CM3(RMS delay = 11 nsec)

A basic training sequence considered for the simulation is a 31 lengthm-sequence padded with 1 to make it an even sequence and a 42 lengthcomplex chirp sequence. Only one basic sequence per sub-band istransmitted to assist timing estimation in a receiver. The comparativeperformance of both of the sequences is demonstrated. Both of thesequences are simulated with the variable frequency diversity orderranging from 1 to 5.

A time-synchronized signal is fed to a de-spreading block afterfrequency domain transformation. After, de-spreading channel estimationis performed by a least square method. The signal is detected afterequalization.

The simulation assumes perfect frequency synchronization. Informationsymbols are spread by an 8 length Walsh-Hadamard code. To supportdiversity, different orthogonal codes are assigned to each sub-band todifferentiate data. The simulation is performed with a single usertransmitting either the same data over a plurality of sub-bands toimprove BER performance (diversity order≧2), or transmitting differentdata over different sub-bands to maximize a data rate. Further, decodingis performed for sub-band 1. To obtain a sampling rate conversion from100 MHz to 500 MHz, the training sequence is up sampled, whereas a datasection is interpolated.

The performance of the 100 MHz synchronizer is compared with theperformance of peak synchronization at 1 GHz to analyze the estimationloss due to sub-sampled estimation. It is observed that for thediversity order of 1, both of the performances match very closely,raising almost zero error in performance. As the diversity orderincreases, a mismatch between 100 MHz and 1 GHz performances alsoincreases. For the zero diversity order, the synchronizer eliminatesdispersion effects due to any one sub-band channel, and aligns anorthogonal code satisfactorily before addition in the method of thede-spreading. The orthogonal code rejects an effect of inter sub-bandinterferences in the process of the de-spreading in this example. Where,for the diversity examples, SNR losses are observed due to loss inorthogonality in between the same codes associated with differentsub-bands. Another cause of this SNR loss contributes to an improperfilter design at terminal filters. As training sequences are designedwith a silence period at a domain, and fine estimation of the timingoffset is performed at a higher sampling rate in analog, it is highlyunlikely that major loss in SNR is contributed by an estimation errorvariance. Hence, the BER performance is verified with up samplinginstead of interpolation for a data section to analyze an effect of apulse shaping filter on the system performance with the same method ofthe timing synchronization.

FIG. 5 is a graph illustrating another example of a comparison of a BERand a SNR of an OFDM S-UWB system with timing synchronization at 1 GHzand 100 MHz over an UWB channel. The BER when a training sequence and adata part of a frame are up-sampled for diversity orders 1 and 5 isshown in FIG. 5. A significant improvement in the SNR of an order of 10dB is observed at the BER of 10⁻³ compared to 1 GHz and 100 MHzperformances with an interpolator for the diversity order of one. TheSNR improvements are even higher for the BERs<10−3. Further, An SNRimprovement of 2 dB is also observed at the same BER compared to 100 MHzperformance with an interpolator for the diversity order of 5. Theperformances of a 100 MHz synchronizer now closely matches with that ofa 1 GHz synchronizer.

FIG. 6 is a diagram illustrating an example of a method of designing atraining sequence for timing synchronization at a sub-sampled bandwidth.The training sequence for a mixed mode timing synchronization method isdescribed herein. The training sequence improves auto correlation andcross correlation properties. Side lobes of a correlation function maybe within a range of 60-70 dB lower than a peak. Further, a correlationproperty of a sequence may be preserved in both a full rate and asub-sampled rate. Further, the training sequence is designed in such away that it can accommodate a latency period of coarse estimation in adigital domain, digital and/or analog circuitry on a feedback path fromdigital to analog, fine estimation in an analog domain, and analogcircuitry generating an equivalent parameter from the fine estimation.

The base training sequence is transmitted at a different timing epoch toestimate related timing offsets of different sub-bands. Further, thebase training sequence occupies a duration of one OFDM symbol. In atotal time period of five OFDM symbols, a training symbol of a sub-bandappears for a duration of one OFDM symbol, and for a rest of the time,it is padded with zeros.

Training sequences for different bands are designed such that they donot overlap with each other at an OFDM symbol duration, over the totalperiod of five OFDM symbols. This ensures that the training sequence foreach sub-band does not overlap with each other even after sub-samplingin a receiver.

In an example, a chirp sequence of length 42 is used as the basicsequence in the design of the training sequence. This complex chirpsequence shows improved auto correlation at a lag zero and near zerocross correlation at other lags.

The training sequence is designed to ensure a provision of a correctestimate of each of these offsets. The design also avoids inter carrierinterference and inter band interference due to timing errors in apractical channel. Furthermore, in view of high data rate transmission,the training sequence is designed such that it poses an efficientbandwidth.

In an example, in order to make a design bandwidth efficient, thetraining sequence for each band is transmitted for one OFDM symbolperiod without any repetition. The accurate offset estimate in eachsub-band is ensured by silence periods for other bands, when thetraining sequence is transmitted for any sub-band. Further, to reduceinter sub-band interferences due to a timing estimation, different typesof a circularly-shifted version of chirp sequences (which confirmsorthogonality up to a shifted delay) may be transmitted for a betterestimate in high delay spread channels.

In an example, the same chirp sequence can be utilized for all of thesub-bands for low delay spread channels. To preserve the correlationproperty of the training sequence in both the full rate and sub-sampleddomains, the basic sequence occupying bandwidth is up-sampled togenerate the sequence in 500 MHz.

FIG. 7 is a schematic diagram illustrating an example of a frame formatof an OFDM-based S-UWB system. Training sequences (preambles) for timingsynchronization are added for a desired band in a preamble sectiondedicated for a frequency synchronization sequence of a frame. Anaccurate offset estimate in each sub-band is ensured by silence periodsfor other bands, when a training sequence is transmitted for anysub-band.

In order to make a design bandwidth efficient the training sequence foreach band is transmitted for one OFDM symbol period without anyrepetition. To reduce an inter sub-band interferences due to timingestimation, a different circularly-shifted version of chirp sequences(which confirms orthogonality up to a shifted delay) may be transmittedfor a better estimate in high delay spread channels.

The training sequences are followed by a silence period of four OFDMsymbol periods as shown in FIG. 7, to accommodate a latency of anelectronic circuit that feeds back a coarse estimate from digital toanalog, a fine estimation, generating of equivalent phase information,and adjusting an ADC clock. However, a duration of the silence periodcan be reduced as the latency of the electronic circuit reduces due toadvancement in device characteristics.

The method and apparatus for timing synchronization described may savepower, cost and chip area. Further, the method and apparatus may achievea good performance, because of a silence period based training sequencedesign.

The various units, blocks, elements, and methods described above may beimplemented using one or more hardware components, one or more softwarecomponents, or a combination of one or more hardware components and oneor more software components.

A hardware component may be, for example, a physical device thatphysically performs one or more operations, but is not limited thereto.Examples of hardware components include microphones, amplifiers,low-pass filters, high-pass filters, band-pass filters,analog-to-digital converters, digital-to-analog converters, andprocessing devices.

A software component may be implemented, for example, by a processingdevice controlled by software or instructions to perform one or moreoperations, but is not limited thereto. A computer, controller, or othercontrol device may cause the processing device to run the software orexecute the instructions. One software component may be implemented byone processing device, or two or more software components may beimplemented by one processing device, or one software component may beimplemented by two or more processing devices, or two or more softwarecomponents may be implemented by two or more processing devices.

A processing device may be implemented using one or more general-purposeor special-purpose computers, such as, for example, a processor, acontroller and an arithmetic logic unit, a digital signal processor, amicrocomputer, a field-programmable array, a programmable logic unit, amicroprocessor, or any other device capable of running software orexecuting instructions. The processing device may run an operatingsystem (OS), and may run one or more software applications that operateunder the OS. The processing device may access, store, manipulate,process, and create data when running the software or executing theinstructions. For simplicity, the singular term “processing device” maybe used in the description, but one of ordinary skill in the art willappreciate that a processing device may include multiple processingelements and multiple types of processing elements. For example, aprocessing device may include one or more processors, or one or moreprocessors and one or more controllers. In addition, differentprocessing configurations are possible, such as parallel processors ormulti-core processors.

A processing device configured to implement a software component toperform an operation A may include a processor programmed to runsoftware or execute instructions to control the processor to performoperation A. In addition, a processing device configured to implement asoftware component to perform an operation A, an operation B, and anoperation C may have various configurations, such as, for example, aprocessor configured to implement a software component to performoperations A, B, and C; a first processor configured to implement asoftware component to perform operation A, and a second processorconfigured to implement a software component to perform operations B andC; a first processor configured to implement a software component toperform operations A and B, and a second processor configured toimplement a software component to perform operation C; a first processorconfigured to implement a software component to perform operation A, asecond processor configured to implement a software component to performoperation B, and a third processor configured to implement a softwarecomponent to perform operation C; a first processor configured toimplement a software component to perform operations A, B, and C, and asecond processor configured to implement a software component to performoperations A, B, and C, or any other configuration of one or moreprocessors each implementing one or more of operations A, B, and C.Although these examples refer to three operations A, B, C, the number ofoperations that may implemented is not limited to three, but may be anynumber of operations required to achieve a desired result or perform adesired task.

Software or instructions for controlling a processing device toimplement a software component may include a computer program, a pieceof code, an instruction, or some combination thereof, for independentlyor collectively instructing or configuring the processing device toperform one or more desired operations. The software or instructions mayinclude machine code that may be directly executed by the processingdevice, such as machine code produced by a compiler, and/or higher-levelcode that may be executed by the processing device using an interpreter.The software or instructions and any associated data, data files, anddata structures may be embodied permanently or temporarily in any typeof machine, component, physical or virtual equipment, computer storagemedium or device, or a propagated signal wave capable of providinginstructions or data to or being interpreted by the processing device.The software or instructions and any associated data, data files, anddata structures also may be distributed over network-coupled computersystems so that the software or instructions and any associated data,data files, and data structures are stored and executed in a distributedfashion.

For example, the software or instructions and any associated data, datafiles, and data structures may be recorded, stored, or fixed in one ormore non-transitory computer-readable storage media. A non-transitorycomputer-readable storage medium may be any data storage device that iscapable of storing the software or instructions and any associated data,data files, and data structures so that they can be read by a computersystem or processing device. Examples of a non-transitorycomputer-readable storage medium include read-only memory (ROM),random-access memory (RAM), flash memory, CD-ROMs, CD-Rs, CD+Rs, CD-RWs,CD+RWs, DVD-ROMs, DVD-Rs, DVD+Rs, DVD-RWs, DVD+RWs, DVD-RAMs, BD-ROMs,BD-Rs, BD-R LTHs, BD-REs, magnetic tapes, floppy disks, magneto-opticaldata storage devices, optical data storage devices, hard disks,solid-state disks, or any other non-transitory computer-readable storagemedium known to one of ordinary skill in the art.

Functional programs, codes, and code segments for implementing theexamples disclosed herein can be easily constructed by a programmerskilled in the art to which the examples pertain based on the drawingsand their corresponding descriptions as provided herein.

While this disclosure includes specific examples, it will be apparent toone of ordinary skill in the art that various changes in form anddetails may be made in these examples without departing from the spiritand scope of the claims and their equivalents. The examples describedherein are to be considered in a descriptive sense only, and not forpurposes of limitation. Descriptions of features or aspects in eachexample are to be considered as being applicable to similar features oraspects in other examples. Suitable results may be achieved if thedescribed techniques are performed in a different order, and/or ifcomponents in a described system, architecture, device, or circuit arecombined in a different manner and/or replaced or supplemented by othercomponents or their equivalents. Therefore, the scope of the disclosureis defined not by the detailed description, but by the claims and theirequivalents, and all variations within the scope of the claims and theirequivalents are to be construed as being included in the disclosure.

What is claimed is:
 1. A method of timing synchronization in sub-bandbased ultra wideband systems, the method comprising: obtaining a coarseestimate of an offset in a time domain at a sub-sampled rate; obtaininga fine estimate of the offset in an analog domain; and correcting atiming in the analog domain by transforming the fine estimate to anequivalent phase for the correcting.
 2. The method as in claim 1,wherein the obtaining of the coarse estimate comprises: performing crosscorrelation of a sub-sampled signal with a sub-sampled training sequencein a digital domain.
 3. The method as in claim 2, wherein the methodfurther comprises: obtaining the coarse estimate for each sub-band basedon a frequency diversity order of a user.
 4. The method as in claim 3,wherein the method further comprises: obtaining a final coarse estimateof the offset from each sub-band by determining a maximum of estimatesfor a corresponding sub-band.
 5. The method as in claim 4, wherein theobtaining of the fine estimate comprises: obtaining the fine estimate inthe analog domain based on the final coarse estimate.
 6. The method asin claim 1, wherein the obtaining of the fine estimate comprises:performing cross correlation of a delayed signal with a trainingsequence designed with a silence period.
 7. The method as in claim 6,wherein a length of the training sequence is equal to a number oforthogonal subcarriers present over a bandwidth of the delayed signal.8. The method as in claim 6, wherein the training sequence is anup-sampled version of a base training sequence.
 9. The method as inclaim 6, wherein the training sequence of a sub-band is orthogonal up toa lag gap of an integer, the lag gap indicating a shift of sub-bands bya value of the integer.
 10. The method as in claim 6, wherein thetraining sequence is followed by the silence period of at least fourorthogonal frequency division multiplexing symbols.
 11. An apparatusconfigured to perform timing synchronization in sub-band based ultrawideband systems, the apparatus comprising: a coarse estimation unitconfigured to obtain a coarse estimate of an offset in a time domain ata sub-sampled rate; a fine estimation unit configured to obtain a fineestimate of the offset in an analog domain; and a timing correction unitconfigured to correct a timing in the analog domain by transforming thefine estimate to an equivalent phase for the correction.
 12. Theapparatus as in claim 11, wherein the coarse estimation unit isconfigured to: perform cross correlation of a sub-sampled signal with asub-sampled training sequence in a digital domain.
 13. The apparatus asin claim 11, wherein the fine estimation unit is configured to: performcross correlation of a delayed signal with a training sequence designedwith a silence period.
 14. The apparatus as in claim 11, furthercomprising: a phase locked loop circuit configured to determine theequivalent phase based on the fine estimate; an analog-to-digitalconverter configured to convert an input signal in the analog domain toa sub-sampled signal in a digital domain to be used to obtain the coarseestimate; and an analog delay circuit configured to delay the inputsignal, the delayed input signal being used to obtain the fine estimate.15. A receiver configured to perform timing synchronization in sub-bandbased ultra wideband systems, the receiver comprising: a processorconfigured to determine a first estimate of a timing offset of an inputsignal, in a digital domain, determine a second estimate of the timingoffset in an analog domain based on the first estimate, and determine anequivalent phase to be used to correct a timing of the input signal,based on the second estimate.
 16. The receiver as in claim 15, whereinthe processor is configured to: sub-sample the input signal to generatea sub-sampled signal in the digital domain; and perform crosscorrelation of the sub-sampled signal with a sub-sampled trainingsequence in the digital domain to generate the first estimate.
 17. Thereceiver as in claim 16, wherein the processor is further configured to:scale the first estimate; and determine the second estimate based on thescaled first estimate.
 18. The receiver as in claim 15, wherein theprocessor is configured to: delay the input signal; and perform crosscorrelation of the delayed input signal with a training sequencefollowed by a silence period, based on the first estimate, to generate acorrelated output.
 19. The receiver as in claim 18, wherein theprocessor is configured to: determine the second estimate to be thefirst estimate in response to the correlated output being greater thanor equal to a predetermined threshold.
 20. The receiver as in claim 15,wherein the processor is configured to: control a switching of ananalog-to-converter clock from an initial phase to the equivalent phasebased on a predetermined timing delay, to correct the timing of theinput signal.